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l t 1110 1 d u escriptio the LT1110 is a versatile micropower dc-dc converter. the device requires only three external components to deliver a fixed output of 5v or 12v. the very low minimum supply voltage of 1.0v allows the use of the LT1110 in applications where the primary power source is a single cell. an on-chip auxiliary gain block can function as a low battery detector or linear post regulator. the 70khz oscillator allows the use of surface mount inductors and capacitors in many applications. quiescent current is just 300 m a, making the device ideal in remote or battery powered applications where current consumption must be kept to a minimum. the device can easily be configured as a step-up or step-down converter, although for most step-down applications or input sources greater than 3v, the lt1111 is recommended. switch current limiting is user-adjustable by adding a single external resistor. unique reverse battery protection circuitry limits reverse current to safe, non- destructive levels at reverse supply voltages up to 1.6v. s f ea t u re n operates at supply voltages from 1.0v to 30v n works in step-up or step-down mode n only three external off-the-shelf components required n low-battery detector comparator on-chip n user-adjustable current limit n internal 1a power switch n fixed or adjustable output voltage versions n space-saving 8-pin minidip or s8 package u s a o pp l ic at i n pagers n cameras n single-cell to 5v converters n battery backup supplies n laptop and palmtop computers n cellular telephones n portable instruments n laser diode drivers n hand-held inventory computers micropower dc-dc converter adjustable and fixed 5v, 12v load current (ma) 0 efficiency (%) 50 60 70 80 85 90 10 20 30 40 LT1110 ?ta02 75 65 55 5152535 v in = 1.50v v in = 1.25v v in = 1.00v u a o pp l ic at i ty p i ca l efficiency all surface mount single cell to 5v converter LT1110 ?ta01 + gnd sw2 sense sw1 lim i in v 54 1 3 8 LT1110-5 1.5v aa cell* *add 10 f decoupling capacitor if battery is more than 2" away from LT1110. m 15 m f tantalum 5v mbrs120t3 sumida cd54-470k 47 m h operates with cell voltage 1.0v 3 2
LT1110 2 wu u package / o rder i for atio a u g w a w u w a r b s o lu t exi t i s supply voltage, step-up mode ................................ 15v supply voltage, step-down mode ........................... 36v sw1 pin voltage ...................................................... 50v sw2 pin voltage ......................................... C 0.5v to v in feedback pin voltage (LT1110) .............................. 5.5v switch current ........................................................ 1.5a maximum power dissipation ............................. 500mw operating temperature range ..................... 0 c to 70 c storage temperature range .................. C65 c to 150 c lead temperature (soldering, 10 sec.)................. 300 c order part number LT1110cn8 LT1110cn8-5 LT1110cn8-12 1110 11105 11012 consult factory for industrial and military grade parts. t jmax = 90 c, q ja = 150 c/w 1 2 3 4 8 7 6 5 top view i lim v in sw1 sw2 fb (sense)* set a0 gnd n8 package 8-lead plastic dip *fixed versions t jmax = 90 c, q ja = 130 c/w 1 2 3 4 8 7 6 5 top view fb (sense)* set a0 gnd i lim v in sw1 sw2 s8 package 8-lead plastic soic *fixed versions symbol parameter conditions min typ max units i q quiescent current switch off l 300 m a v in input voltage step-up mode l 1.15 12.6 v 1.0 12.6 v step-down mode l 30 v comparator trip point voltage LT1110 (note 1) l 210 220 230 mv v out output sense voltage LT1110-5 (note 2) l 4.75 5.00 5.25 v LT1110-12 (note 2) l 11.4 12.00 12.6 v comparator hysteresis LT1110 l 48 mv output hysteresis LT1110-5 l 90 180 mv LT1110-12 l 200 400 mv f osc oscillator frequency l 52 70 90 khz dc duty cycle full load (v fb < v ref ) l 62 69 78 % t on switch on time l 7.5 10 12.5 m s i fb feedback pin bias current LT1110, v fb = 0v l 70 150 na i set set pin bias current v set = v ref l 100 300 na v ao ao output low i ao = C300 m a, v set = 150mv l 0.15 0.4 v reference line regulation 1.0v v in 1.5v l 0.35 1.0 %/v 1.5v v in 12v l 0.05 0.1 %/v e lectr ic al c c hara terist ics t a = 25 c, v in = 1.5v, unless otherwise noted. s8 part marking LT1110cs8 LT1110cs8-5 LT1110cs8-12
l t 1110 3 v cesat switch saturation voltage v in = 1.5v, i sw = 400ma 300 400 mv step-up mode l 600 mv v in = 1.5v, i sw = 500ma 400 550 mv l 750 mv v in = 5v, i sw = 1a 700 1000 mv a v a2 error amp gain r l = 100k w (note 3) l 1000 5000 v/v i rev reverse battery current (note 4) 750 ma i lim current limit 220 w between i lim and v in 400 ma current limit temperature C 0.3 %/ c coefficient i leak switch off leakage current measured at sw1 pin 1 10 m a v sw2 maximum excursion below gnd i sw1 10 m a, switch off C 400 C 350 mv e lectr ic al c c hara terist ics t a = 25 c, v in = 1.5v, unless otherwise noted. symbol parameter conditions min typ max units note 3: 100k w resistor connected between a 5v source and the ao pin. note 4: the LT1110 is guaranteed to withstand continuous application of +1.6v applied to the gnd and sw2 pins while v in , i lim , and sw1 pins are grounded. the l denotes the specifications which apply over the full operating temperature range. note 1: this specification guarantees that both the high and low trip point of the comparator fall within the 210mv to 230mv range. note 2: this specification guarantees that the output voltage of the fixed versions will always fall within the specified range. the waveform at the sense pin will exhibit a sawtooth shape due to the comparator hysteresis. cc hara terist ics uw a t y p i ca lper f o r c e oscillator frequency oscillator frequency switch on time input voltage (v) 0 64 frequency (khz) 66 70 72 74 78 80 36 912 LT1110 ?tpc02 15 18 21 62 60 24 27 30 68 76 temperature ( c) on time ( m s) 50 ?5 0 25 LT1110 ?tpc03 50 75 100 14 13 12 11 10 9 8 7 temperature ( c) ?0 40 oscillator frequency (khz) 50 60 70 80 90 100 ?5 0 25 50 LT1110 ?tpc01 75 100
LT1110 4 i switch (a) on voltage (v) 0 0.2 0.4 0.6 LT1110 ?tpc07 0.8 1.0 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 v in = 12v cc hara terist ics uw a t y p i ca lper f o r c e saturation voltage duty cycle switch saturation voltage step-up mode switch on voltage minimum/maximum frequency vs step-down mode on time quiescent current maximum switch current vs maximum switch current vs quiescent current r lim step-up r lim step-down temperature ( c) duty cycle (%) 50 ?5 0 25 LT1110 ?tpc04 50 75 100 78 76 74 72 70 68 66 64 62 60 58 temperature ( c) v cesat (mv) 50 25 0 25 LT1110 ?tpc05 50 75 100 500 450 400 350 300 250 200 150 100 50 0 v in = 1.5v i sw = 500ma i (a) 0 0 v (v) 0.2 0.4 0.6 1.2 1.4 0.2 0.4 0.8 1.2 LT1110 ?tpc06 1.0 1.4 1.6 switch cesat v in = 1.0v v = 1.2v in v in = 1.5v v in = 5.0v v = 2.0v in 0.6 1.0 0.8 v in = 3.0v input voltage (v) quiescent current ( m a) 0 LT1110 ?tpc09 3 400 380 360 340 320 280 260 6 240 220 200 300 9 12151821242730 temperature ( c) quiescent current ( m a) ?0 LT1110 ?tpc10 500 450 400 350 250 ?5 150 100 200 300 0 25 50 75 100 r lim ( w ) switch current (a) 10 LT1110 ?tpc11 100 1.5 1.3 1.1 0.9 1000 0.7 0.5 0.3 0.1 step-up mode v in 5v r lim ( w ) switch current (a) 10 LT1110 ?tpc12 100 1.5 1.3 1.1 0.9 1000 0.7 0.5 0.3 0.1 step-down mode v in = 12v switch on time ( m s) oscillator frequency (khz) 7 9 LT1110 ?tpc08 10 100 90 80 70 60 50 40 13 0 c t a 70 c 8 11 12 95 85 75 65 55 45
l t 1110 5 w i dagra b l o c k lt 1110 cc hara terist ics uw a t y p i ca lper f o r c e set pin bias current fb pin bias current reference voltage i lim (pin 1): connect this pin to v in for normal use. where lower current limit is desired, connect a resistor between i lim and v in . a 220 w resistor will limit the switch current to approximately 400ma. v in (pin 2): input supply voltage. sw1 (pin 3): collector of power transistor. for step-up mode connect to inductor/diode. for step-down mode connect to v in . sw2 (pin 4): emitter of power transistor. for step-up mode connect to ground. for step-down mode connect to inductor/diode. this pin must never be allowed to go more than a schottky diode drop below ground. gnd (pin 5): ground. ao (pin 6): auxiliary gain block (gb) output. open collector, can sink 300 m a. set (pin 7): gb input. gb is an op amp with positive input connected to set pin and negative input connected to 220mv reference. fb/sense (pin 8): on the LT1110 (adjustable) this pin goes to the comparator input. on the LT1110-5 and LT1110-12, this pin goes to the internal application resistor that sets output voltage. pi u fu u c u s o ti temperature ( c) bias current (na) ?0 LT1110 ?tpc13 160 140 120 100 60 ?5 20 0 40 80 0 25 50 75 100 temperature ( c) v ref (mv) 50 ?5 0 25 LT1110 ?tpc15 50 75 100 226 224 222 220 218 216 214 212 temperature ( c) bias current (na) ?0 120 100 90 70 40 ?5 10 0 30 60 0 25 50 75 100 20 50 80 110 LT1110 ?tpc14 LT1110 ?bd01 in v gnd set ao gain block/error amp 220mv reference a1 a2 driver + fb sw1 sw2 lim i oscillator comparator q1
LT1110 6 - lt 1110 u oper o at i lt 1110 the LT1110 is a gated oscillator switcher. this type architecture has very low supply current because the switch is cycled only when the feedback pin voltage drops below the reference voltage. circuit operation can best be understood by referring to the LT1110 block diagram above. comparator a1 compares the fb pin voltage with the 220mv reference signal. when fb drops below 220mv, a1 switches on the 70khz oscillator. the driver amplifier boosts the signal level to drive the output npn power switch q1. an adaptive base drive circuit senses switch current and provides just enough base drive to ensure switch saturation without overdriving the switch, resulting in higher efficiency. the switch cycling action raises the output voltage and fb pin voltage. when the fb voltage is sufficient to trip a1, the oscillator is gated off. a small amount of hysteresis built into a1 ensures loop stability without external frequency compensation. when the comparator is low the oscillator and all high current circuitry is turned off, lowering device quiescent current to just 300 m a for the reference, a1 and a2. the oscillator is set internally for 10 m s on time and 5 m s off time, optimizing the device for step-up circuits where v out ? 3v in , e.g., 1.5v to 5v. other step-up ratios as well as step-down (buck) converters are possible at slight losses in maximum achievable power output. a2 is a versatile gain block that can serve as a low battery detector, a linear post regulator, or drive an under voltage lockout circuit. the negative input of a2 is internally connected to the 220mv reference. an external resistor divider from v in to gnd provides the trip point for a2. the ao output can sink 300 m a (use a 47k resistor pull up to +5v). this line can signal a microcontroller that the battery voltage has dropped below the preset level. to prevent the gain block from operating in its linear region, a 2m w resistor can be connected from ao to set. this provides positive feedback. a resistor connected between the i lim pin and v in adjusts maximum switch current. when the switch current ex- ceeds the set value, the switch is turned off. this feature is especially useful when small inductance values are used with high input voltages. if the internal current limit of 1.5a is desired, i lim should be tied directly to v in . propagation delay through the current limit circuitry is about 700ns. in step-up mode, sw2 is connected to ground and sw1 drives the inductor. in step-down mode, sw1 is con- nected to v in and sw2 drives the inductor. output voltage is set by the following equation in either step-up or step- down modes where r1 is connected from fb to gnd and r2 is connected from v out to fb. the LT1110-5 and LT1110-12 fixed output voltage ver- sions have the gain setting resistors on-chip. only three external components are required to construct a 5v or 12v output converter. 16 m a flows through r1 and r2 in the LT1110-5, and 39 m a flows in the LT1110-12. this current represents a load and the converter must cycle from time to time to maintain the proper output voltage. output ripple, inherently present in gated oscillator designs, will typically run around 90mv for the LT1110-5 and 200mv for the LT1110-12 with the proper inductor/capacitor selection. this output ripple can be reduced considerably by using the gain block amp as a pre-amplifier in front of the fb pin. see the applications section for details. w i dagra b l o c k -5, -12 lt 1110 a1 LT1110 ?bd02 in v gnd set ao a2 220mv ref oscillator driver + r1 sw1 sw2 lim i r2 300k w sense LT1110-5: LT1110-12: r1 = 13.8k w r2 = 5.6k w gain block/error amp comparator q1 u oper o at i -5, -12 vmv r r out = () + ? ? ? ? 220 2 1 101 .()
l t 1110 7 inductor selection general a dc-dc converter operates by storing energy as mag- netic flux in an inductor core, and then switching this energy into the load. since it is flux, not charge, that is stored, the output voltage can be higher, lower, or oppo- site in polarity to the input voltage by choosing an appro- priate switching topology. to operate as an efficient en- ergy transfer element, the inductor must fulfill three re- quirements. first, the inductance must be low enough for the inductor to store adequate energy under the worst case condition of minimum input voltage and switch on time. the inductance must also be high enough so maxi- mum current ratings of the LT1110 and inductor are not exceeded at the other worst case condition of maximum input voltage and on time. additionally, the inductor core must be able to store the required flux; i.e., it must not saturate . at power levels generally encountered with LT1110 based designs, small surface mount ferrite core units with saturation current ratings in the 300ma to 1a range and dcr less than 0.4 w (depending on application) are adequate. lastly, the inductor must have sufficiently low dc resistance so excessive power is not lost as heat in the windings. an additional consideration is electro- magnetic interference (emi). toroid and pot core type inductors are recommended in applications where emi must be kept to a minimum; for example, where there are sensitive analog circuitry or transducers nearby. rod core types are a less expensive choice where emi is not a problem. minimum and maximum input voltage, output voltage and output current must be established before an inductor can be selected. inductor selection step-up converter in a step-up, or boost converter (figure 4), power gener- ated by the inductor makes up the difference between input and output. power required from the inductor is determined by pv vv i l out d in min out =+ ()() () 01 where v d is the diode drop (0.5v for a 1n5818 schottky). u s a o pp l ic at i wu u i for atio energy required by the inductor per cycle must be equal or greater than p f l osc () 02 in order for the converter to regulate the output. when the switch is closed, current in the inductor builds according to it v r e l in rt l () ? () ? = ? ? ? ? 103 where r' is the sum of the switch equivalent resistance (0.8 w typical at 25 c) and the inductor dc resistance. when the drop across the switch is small compared to v in , the simple lossless equation it v l t l in () = () 04 can be used. these equations assume that at t = 0, inductor current is zero. this situation is called discon- tinuous mode operation in switching regulator parlance. setting t to the switch on time from the LT1110 speci- fication table (typically 10 m s) will yield i peak for a specific l and v in . once i peak is known, energy in the inductor at the end of the switch on time can be calculated as eli l peak = 1 2 05 2 () e l must be greater than p l /f osc for the converter to deliver the required power. for best efficiency i peak should be kept to 1a or less. higher switch currents will cause excessive drop across the switch resulting in reduced efficiency. in general, switch current should be held to as low a value as possible in order to keep switch, diode and inductor losses at a minimum. as an example, suppose 12v at 120ma is to be generated from a 4.5v to 8v input. recalling equation (01), p v v v ma mw l =+ ()() = 12 0 5 4 5 120 960 06 .. .() energy required from the inductor is p f mw khz j l osc == 960 70 13 7 07 .. () m
LT1110 8 u s a o pp l ic at i wu u i for atio picking an inductor value of 47 m h with 0.2 w dcr results in a peak switch current of i v ema peak s h =- ? ? = - 45 10 1 862 08 10 10 47 . . .() . w wm m substituting i peak into equation 05 results in ehaj l = ()( ) = 1 2 47 0 862 17 5 09 2 mm ...() since 17.5 m j > 13.7 m j, the 47 m h inductor will work. this trial-and-error approach can be used to select the opti- mum inductor. keep in mind the switch current maximum rating of 1.5a. if the calculated peak current exceeds this, an external power transistor can be used. a resistor can be added in series with the i lim pin to invoke switch current limit. the resistor should be picked such that the calculated i peak at minimum v in is equal to the maximum switch current (from typical performance characteristic curves). then, as v in increases, switch current is held constant, resulting in increasing efficiency. inductor selection step-down converter the step-down case (figure 5) differs from the step-up in that the inductor current flows through the load during both the charge and discharge periods of the inductor. current through the switch should be limited to ~800ma in this mode. higher current can be obtained by using an external switch (see figure 6). the i lim pin is the key to successful operation over varying inputs. after establishing output voltage, output current and input voltage range, peak switch current can be calculated by the formula i i dc vv vv v peak out out d in sw d = + + ? ? 2 10 () where dc = duty cycle (0.69) v sw = switch drop in step-down mode v d = diode drop (0.5v for a 1n5818) i out = output current v out = output voltage v in = minimum input voltage v sw is actually a function of switch current which is in turn a function of v in , l, time and v out . to simplify, 1.5v can be used for v sw as a very conservative value. once i peak is known, inductor value can be derived from l vvv i t in min sw out peak on = () 11 where t on = switch on time (10 m s). next, the current limit resistor r lim is selected to give i peak from the r lim step-down mode curve. the addition of this resistor keeps maximum switch current constant as the input voltage is increased. as an example, suppose 5v at 250ma is to be generated from a 9v to 18v input. recalling equation (10), i ma ma peak = () + + ? ? = 2 250 069 505 91505 498 12 . . . . .( ) next, inductor value is calculated using equation (11) l ma sh == 9155 498 10 50 13 . .() mm use the next lowest standard value (47 m h). then pick r lim from the curve. for i peak = 500ma, r lim = 82 w . inductor selection positive-to-negative converter figure 7 shows hookup for positive-to-negative conver- sion. all of the output power must come from the inductor. in this case, pv vi l out d out =+ ()() || . () 14 in this mode the switch is arranged in common collector or step-down mode. the switch drop can be modeled as a 0.75v source in series with a 0.65 w resistor. when the
l t 1110 9 switch closes, current in the inductor builds according to i v r e l l rt l + () = ? ? ? ? ? () ? 115 where r' = 0.65 w + dcr l v l = v in C 0.75v as an example, suppose C5v at 75ma is to be generated from a 4.5v to 5.5v input. recalling equation (14), pvvmamw l =- + ()() = || ..() 5 0 5 75 413 16 energy required from the inductor is p f mw khz j l osc == 413 70 59 17 .. () m picking an inductor value of 56 m h with 0.2 w dcr results in a peak switch current of i vv ema peak s h = () + () ? ? ? ? = 45 075 065 02 1 621 18 085 10 56 .. .. .() . ww wm m substituting i peak into equation (04) results in ehaj l = ()( ) = 1 2 56 0 621 10 8 19 2 mm ...() since 10.8 m j > 5.9 m j, the 56 m h inductor will work. with this relatively small input range, r lim is not usually necessary and the i lim pin can be tied directly to v in . as in the step-down case, peak switch current should be limited to ~800ma. capacitor selection selecting the right output capacitor is almost as important as selecting the right inductor. a poor choice for a filter capacitor can result in poor efficiency and/or high output ripple. ordinary aluminum electrolytics, while inexpensive and readily available, may have unacceptably poor equiva- lent series resistance (esr) and esl (inductance). there are low esr aluminum capacitors on the market specifi- cally designed for switch mode dc-dc converters which work much better than general-purpose units. tantalum capacitors provide still better performance at more ex- pense. we recommend os-con capacitors from sanyo corporation (san diego, ca). these units are physically quite small and have extremely low esr. to illustrate, figures 1, 2 and 3 show the output voltage of an LT1110 based converter with three 100 m f capacitors. the peak switch current is 500ma in all cases. figure 1 shows a sprague 501d, 25v aluminum capacitor. v out jumps by over 120mv when the switch turns off, followed by a drop in voltage as the inductor dumps into the capacitor. this works out to be an esr of over 240m w . figure 2 shows the same circuit, but with a sprague 150d, 20v tantalum capacitor replacing the aluminum unit. output jump is now about 35mv, corresponding to an esr of 70m w . figure 3 shows the circuit with a 16v os-con unit. esr is now only 20m w . 5 s/div 50mv/div LT1110 ?ta19 m figure 1. aluminum 5 s/div 50mv/div LT1110 ?ta20 m figure 2. tantalum 5 s/div 50mv/div LT1110 ?ta21 m figure 3. os-con u s a o pp l ic at i wu u i for atio
LT1110 10 diode selection speed, forward drop, and leakage current are the three main considerations in selecting a catch diode for LT1110 converters. general purpose rectifiers such as the 1n4001 are unsuitable for use in any switching regulator applica- tion. although they are rated at 1a, the switching time of a 1n4001 is in the 10 m s-50 m s range. at best, efficiency will be severely compromised when these diodes are used; at worst, the circuit may not work at all. most LT1110 circuits will be well served by a 1n5818 schottky diode, or its surface mount equivalent, the mbrs130t3. the combina- tion of 500mv forward drop at 1a current, fast turn on and turn off time, and 4 m a to 10 m a leakage current fit nicely with LT1110 requirements. at peak switch currents of 100ma or less, a 1n4148 signal diode may be used. this diode has leakage current in the 1na-5na range at 25 c and lower cost than a 1n5818. (you can also use them to get your circuit up and running, but beware of destroying the diode at 1a switch currents.) step-up (boost mode) operation a step-up dc-dc converter delivers an output voltage higher than the input voltage. step-up converters are not short circuit protected since there is a dc path from input to output. the usual step-up configuration for the LT1110 is shown in figure 4. the LT1110 first pulls sw1 low causing v in C v cesat to appear across l1. a current then builds up in l1. at the end of the switch on time the current in l1 is 1 : i v l t pea k in on = () 20 u s a o pp l ic at i wu u i for atio immediately after switch turn off, the sw1 voltage pin starts to rise because current cannot instantaneously stop flowing in l1. when the voltage reaches v out + v d , the inductor current flows through d1 into c1, increasing v out . this action is repeated as needed by the LT1110 to keep v fb at the internal reference voltage of 220mv. r1 and r2 set the output voltage according to the formula v r r mv out =+ ? ? ? ? () 1 2 1 220 21 .() step-down (buck mode) operation a step-down dc-dc converter converts a higher voltage to a lower voltage. the usual hookup for an LT1110 based step-down converter is shown in figure 5. LT1110 ?ta15 gnd sw2 sw1 lim i in v r3 220 fb v out + c2 + c1 d1 1n5818 v in r2 r1 l1 w LT1110 figure 5. step-down mode hookup when the switch turns on, sw2 pulls up to v in C v sw . this puts a voltage across l1 equal to v in C v sw C v out , causing a current to build up in l1. at the end of the switch on time, the current in l1 is equal to i v vv l t peak in sw out on = -- .() 22 when the switch turns off, the sw2 pin falls rapidly and actually goes below ground. d1 turns on when sw2 reaches 0.4v below ground. d1 must be a schottky diode . the voltage at sw2 must never be allowed to go below C0.5v. a silicon diode such as the 1n4933 will allow sw2 to go to C0.8v, causing potentially destructive power figure 4. step-up mode hookup. l1 LT1110 ?ta14 gnd sw2 sw1 lim i in v d1 r3* LT1110 + v out r2 r1 c1 * = optional v in fb note 1 : this simple expression neglects the effects of switch and coil resistance. this is taken into account in the inductor selection section.
l t 1110 11 converter section with the following conservative ex- pression for v sw : vvv v sw r sat =+ ? 1 09 24 .. () r2 provides a current path to turn off q1. r3 provides base drive to q1. r4 and r5 set output voltage. inverting configurations the LT1110 can be configured as a positive-to-negative converter (figure 7), or a negative-to-positive converter (figure 8). in figure 7, the arrangement is very similar to a step-down, except that the high side of the feedback is referred to ground. this level shifts the output negative. as in the step-down mode, d1 must be a schottky diode, and ? v out ? should be less than 6.2v. more negative out- put voltages can be accommodated as in the prior section. LT1110 ?ta03 ? out + c2 + c1 d1 1n5818 +v in r1 r2 l1 gnd sw2 sw1 lim i in v r3 fb LT1110 figure 7. positive-to-negative converter in figure 8, the input is negative while the output is positive. in this configuration, the magnitude of the input voltage can be higher or lower than the output voltage. a level shift, provided by the pnp transistor, supplies proper polarity feedback information to the regulator. dissipation inside the LT1110. output voltage is deter- mined by v r r mv out =+ ? ? ? ? () 1 2 1 220 23 .() r3 programs switch current limit. this is especially im- portant in applications where the input varies over a wide range. without r3, the switch stays on for a fixed time each cycle. under certain conditions the current in l1 can build up to excessive levels, exceeding the switch rating and/or saturating the inductor. the 220 w resistor pro- grams the switch to turn off when the current reaches approximately 800ma. when using the LT1110 in step- down mode, output voltage should be limited to 6.2v or less. higher output voltages can be accommodated by inserting a 1n5818 diode in series with the sw2 pin (anode connected to sw2). higher current step-down operation output current can be increased by using a discrete pnp pass transistor as shown in figure 6. r1 serves as a current limit sense. when the voltage drop across r1 equals a v be , the switch turns off. for temperature com- pensation a schottky diode can be inserted in series with the i lim pin. this also lowers the maximum drop across r1 to v be C v d , increasing efficiency. as shown, switch current is limited to 2a. inductor value can be calculated based on formulas in the inductor selection step-down u s a o pp l ic at i wu u i for atio figure 6. q1 permits higher-current switching. LT1110 functions as controller. LT1110 ?ta16 d1 1n5821 + + v out v in 25v max l1 r1 0.3 w r2 220 q1 mje210 or zetex ztx789a r3 330 r4 r5 c1 v out = 220mv ( 1 + ) r4 r5 LT1110 gnd sw2 sw1 v in i l fb c2 figure 8. negative-to-positive converter l1 LT1110 ?ta04 gnd sw2 fb sw1 lim i in v d1 ao +v out r2 v out = 220mv + 0.6v r1 r2 ( ) r1 2n3906 ? in + c1 LT1110 + c2
LT1110 12 u s a o pp l ic at i wu u i for atio using the i lim pin the LT1110 switch can be programmed to turn off at a set switch current, a feature not found on competing devices. this enables the input to vary over a wide range without exceeding the maximum switch rating or saturating the inductor. consider the case where analysis shows the LT1110 must operate at an 800ma peak switch current with a 2.0v input. if v in rises to 4v, peak current will rise to 1.6a, exceeding the maximum switch current rating. with the proper resistor selected (see the maximum switch current vs r lim characteristic), the switch current will be limited to 800ma, even if the input voltage increases. another situation where the i lim feature is useful occurs when the device goes into continuous mode operation. this occurs in step-up mode when v v vv dc out diode in sw + - < - 1 1 25 .() when the input and output voltages satisfy this relation- ship, inductor current does not go to zero during the switch off time. when the switch turns on again, the current ramp starts from the non-zero current level in the inductor just prior to switch turn on. as shown in figure 9, the inductor current increases to a high level before the comparator turns off the oscillator. this high current can cause excessive output ripple and requires oversizing the output capacitor and inductor. with the i lim feature, however, the switch current turns off at a programmed level as shown in figure 10, keeping output ripple to a minimum. figure 11 details current limit circuitry. sense transistor q1, whose base and emitter are paralleled with power switch q2, is ratioed such that approximately 0.5% of q2s collector current flows in q1s collector. this current is passed through internal 80 w resistor r1 and out through the i lim pin. the value of the external resistor connected between i lim and v in set the current limit. when sufficient switch current flows to develop a v be across r1 + r lim , q3 turns on and injects current into the oscillator, turning off the switch. delay through this circuitry is approximately 800ns. the current trip point becomes less accurate for switch on times less than 3 m s. resistor values program- ming switch on time for 800ns or less will cause spurious response in the switch circuitry although the device will still maintain output regulation. LT1110 ?ta05 i off l on switch figure 9. no current limit causes large inductor current build-up LT1110 ?ta06 i on l off switch programmed current limit figure 10. current limit keeps inductor current under control LT1110 ?ta17 sw2 sw1 q2 driver oscillator v in i lim r1 80 w (internal) r lim (external) q1 q3 figure 11. LT1110 current limit circuitry using the gain block the gain block (gb) on the LT1110 can be used as an error amplifier, low battery detector or linear post regulator. the gain block itself is a very simple pnp input op amp with an open collector npn output. the negative input of the gain block is tied internally to the 220mv reference. the posi- tive input comes out on the set pin.
l t 1110 13 arrangement of the gain block as a low battery detector is straightforward. figure 12 shows hookup. r1 and r2 need only be low enough in value so that the bias current of the set input does not cause large errors. 33k w for r2 is adequate. r3 can be added to introduce a small amount of hysteresis. this will cause the gain block to snap when the trip point is reached. values in the 1m-10m range are optimal. the addition of r3 will change the trip point, however. u s a o pp l ic at i wu u i for atio output ripple of the LT1110, normally 90mv at 5v out can be reduced significantly by placing the gain block in front of the fb input as shown in figure 13. this effectively reduces the comparator hysteresis by the gain of the gain block. output ripple can be reduced to just a few millivolts using this technique. ripple reduction works with step- down or inverting modes as well. for this technique to be effective, output capacitor c1 must be large, so that each switching cycle increases v out by only a few millivolts. 1000 m f is a good starting value. figure 13. output ripple reduction using gain block figure 12. setting low battery detector trip point LT1110 ?ta07 v bat r1 r2 220mv ref set gnd v in LT1110 47k +5v to processor r1 = v lb ?220mv ( ) 4.33 m a v lb = battery trip point + ao r3 r2 = 33k w r3 = 2m w l1 LT1110 ?ta08 gnd sw2 set sw1 lim i in v d1 r3 270k fb + v out r2 r1 c1 v = + 1 220mv out r2 r1 ( ) ( ) LT1110 ao v bat table 2. capacitor manufacturers manufacturer part numbers sanyo video components os-con series 2001 sanyo avenue san diego, ca 92173 619-661-6835 nichicon america corporation pl series 927 east state parkway schaumberg, il 60173 708-843-7500 sprague electric company 150d solid tantalums lower main street 550d tantalex sanford, me 04073 207-324-4140 matsuo 267 series 714-969-2491 surface mount table 3. transistor manufacturers manufacturer part numbers zetex ztx series commack, ny fzt series 516-543-7100 surface mount table 1. inductor manufacturers manufacturer part numbers coiltronics international ctx100-4 series 984 s.w. 13th court surface mount pompano beach, fl 33069 305-781-8900 sumida electric co. usa cd54 708-956-0666 cdr74 cdr105 surface mount
LT1110 14 u s a o pp l ic at i ty p i ca l all surface mount flash memory v pp generator 1.5v powered laser diode driver l1 2.2 h LT1110 ?ta13 m c1 100 f os-con m 1n5818 4.7k 10 1k* r1 + toshiba told-9211 w 220 w 2n3906 12 3 7 4 5 6 8 22nf 1.5v adjust r1 for change in laser output power toko 262lyf-0076m * 1n4148 2 w gnd sw2 set sw1 lim i in v fb ao LT1110 laser diode case common to + battery terminal 170ma current drain from 1.5v cell (50ma diode) no overshoot ? ? ? = = mje210 0.22 f ceramic m 1.5v powered laser diode driver LT1110 ?ta18 gnd sw2 sense sw1 lim i in v LT1110cs8-12 *l1= sumida cd105-470m 47 m f 20v l1* 47 m h v pp 12v 120ma 22 m f 10k +5v ?0% + + 1k mbrs12ot3 mmbt4403 = program = shutdown 1 0 mmbf170
l t 1110 15 LT1110 ?ta10 gnd sw2 sense sw1 lim i in v LT1110-5 9v *l1 = coilcraft 1812ls-473 10 m f mbrl120 5v 40ma 220 l1* 47 m h + LT1110 ?ta11 gnd sw2 fb sw1 lim i in v LT1110 1.5v aa or aaa cell = mbrl120 = coilcraft 1812ls-823 4.7 m f l1* 82 m h +5v 3ma 4.7 m f +10v 3ma 490k *l1 4.7 m f 11k + + + LT1110 ?ta09 gnd sw2 sense sw1 lim i in v LT1110-5 3v 2x aa cell *l1 = coilcraft 1812ls-473 10 m f mbrl120 l1* 47 m h 5v 40ma 220 + LT1110 ?ta12 gnd sw2 sense sw1 lim i in v LT1110 1.5v aa or aaa cell = mbrl120 = coilcraft 1812ls-823 4.7 m f l1* 82 m h +5v 4ma 4.7 m f ?v 4ma 4.7 m f + + *l1 + u s a o pp l ic at i ty p i ca l all surface mount 3v to 5v step-up converter all surface mount 9v to 5v step-down converter all surface mount 1.5v to +10v, +5v dual output step-up converter all surface mount 1.5v to 5v dual output step-up converter information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LT1110 16 u package d e sc r i pti o dimensions in inches (millimeters) unless otherwise noted. n8 package 8-lead plastic dip s8 package 8-lead plastic soic ? linear technology corporation 1994 linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7487 (408) 432-1900 l fax : (408) 434-0507 l telex : 499-3977 lt/gp 0594 2k rev b ? printed in usa 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 8?typ 0.008 ?0.010 (0.203 ?0.254) 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157* (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.006 inch (0.15mm). 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.320 (7.620 ?8.128) 0.325 +0.025 0.015 +0.635 0.381 8.255 () 0.045 ?0.015 (1.143 ?0.381) 0.100 ?0.010 (2.540 ?0.254) 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 ?0.005 (3.302 ?0.127) 0.020 (0.508) min 0.018 ?0.003 (0.457 ?0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.250 ?0.010 (6.350 ?0.254) 0.400 (10.160) max

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